Benchtop Power Supply Options

[whit3rd]

On Wednesday, May 28, 2014 7:27:09 PM UTC-7, Phil Hobbs wrote:

On 5/28/2014 4:20 PM, whit3rd wrote:

Have you considered direct shunt regulation?
[ AC-coupled example, with some DC stability issues unaddressed]

You're stuck with the wideband noise of the reference, though. I often
have to care about nanovolt 1-Hz noise on power supplies.

Classic solution: a string of mercury batteries. You can still do it, with
standard cells.

Wideband noise in a '431 is likely recombination noise in the base currents,
as I understand it. So, maybe one with higher quiescent current would
be better. A FET solution (like current-limit diode for a reference and
jFET op amp for gain) would help, if things like popcorn/flicker didn't
take over.

Even the humble follower transistor in the lightly-loaded C-multiplier, might
be noisier than a FET (or MOSFET,though I recall a discussion that showed
the MOSFET to have no advantage at Zgate under 100k ohms).

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.
Maybe the heatpump items aren't optimized for it, but one could
imagine using two-alloy printed wiring to get to some serious
potential differences in a compact, rugged, ultrareliable component.

You'd just have to allow a bit of warm-up time on that photon detector.

 

[Phil Hobbs]

On 05/28/2014 01:15 PM, John Larkin wrote:

On Wed, 28 May 2014 13:05:31 -0400, Phil Hobbs
pcdhSpamMeSenseless@electrooptical.net> wrote:

On 05/28/2014 10:24 AM, George Herold wrote:
On Tuesday, May 27, 2014 8:27:19 PM UTC-4, Phil Hobbs wrote:
On 5/27/2014 7:21 PM, John Larkin wrote:

On Mon, 26 May 2014 21:50:17 -0400, Phil Hobbs


Lately I've been using shunt regulators on the base string of cap
multipliers. AFAICT the best is the Exar SPX431A. You can adjust that
to lop off the ripple if you like, though of course it isn't as
efficient as using a huge cap.

That's interesting, the Exar is "best" because it has the least noise to start with?

As you've said, the reference/regulator can be the noisiest part of a design.
I've got an LT3080 in an instrument. (cap multiplier after it.)
And I've regretted it. Much better to just make your own,
(reference->opamp->transistor)

George H.


The Exar is quieter and has a lower minimum cathode current.

0-------*---------RRRR---*---* *-*----------0
| | \ / |
R | \ A |
R | ------ |
R CCC | |
R CCC R |
| | R R
| GND R R
| R R
| | R
| | |
*---RRRR---*--RRRR------* |
| | | |
| | | |
CCC /---/ CCC |
CCC / \---* CCC *--RRRR--*
| --- | | | |
GND | | GND | GND
| | |
GND *-------------*

Cheers

Phil Hobbs

That circuit necessarily increases Vce, which has additional benefits.

Lots of circuits have good supply rejection at low frequencies and
need help at high frequencies. In those cases, just RC power filtering
works well. Polymer aluminum caps are great for that.

I've also done closed-loop opamp regulators with a huge RC tau on the
output. That lops off the HF part of the noise.

Yup. A two-pole cap multiplier (the above circuit with another RC stage
in front of the regulator) can do 100-dB of suppression at SMPS
frequencies, which is what I usually want it for.

Connecting the cold end of a photodiode to a supply that isn't
super-quiet is one good way to blow the signal out of the water.

Cheers

Phil Hobbs


--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[Phil Hobbs]

On 05/28/2014 04:20 PM, whit3rd wrote:

On Wednesday, May 28, 2014 10:05:31 AM UTC-7, Phil Hobbs wrote:

Lately I've been using shunt regulators on the base string of cap

multipliers. AFAICT the best is the Exar SPX431A.
The Exar is quieter and has a lower minimum cathode current.

It's possible to get lower (SPX431A wants over 400 uA, and TLV431
wants over 50 uA) but there's some cathode-voltage-range issues, too.

Have you considered direct shunt regulation?



0--------*---RRRR---*-----*----------0
| | |
R | |
R | *-----*
R CCC R |
R CCC R C
| | R |
| | *-----*
*--RRRR----* |
| | |
| /---/ |
CCC / \----*
CCC --- |
| | R
| | R
GND | |
GND GND

That only helps within the regulator's bandwidth, and the maximum
suppression is limited to its loop gain.

The cap multiplier is quiet at all frequencies of interest (usually),
and the regulator just keeps its output from wandering around.

For running op amps, a barefoot cap multiplier is usually better,
because it's simpler and quieter, very stiff at high frequency, and you
usually don't care about a few hundred millivolts' worth of supply sag.


Cheers

Phil Hobbs
--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[George Herold]

On Thursday, May 29, 2014 1:44:41 PM UTC-4, whit3rd wrote:

On Wednesday, May 28, 2014 7:27:09 PM UTC-7, Phil Hobbs wrote:

On 5/28/2014 4:20 PM, whit3rd wrote:



Have you considered direct shunt regulation?

[ AC-coupled example, with some DC stability issues unaddressed]



You're stuck with the wideband noise of the reference, though. I often

have to care about nanovolt 1-Hz noise on power supplies.



Classic solution: a string of mercury batteries. You can still do it, with

standard cells.



Wideband noise in a '431 is likely recombination noise in the base currents,

as I understand it. So, maybe one with higher quiescent current would

be better. A FET solution (like current-limit diode for a reference and

jFET op amp for gain) would help, if things like popcorn/flicker didn't

take over.



Even the humble follower transistor in the lightly-loaded C-multiplier, might

be noisier than a FET (or MOSFET,though I recall a discussion that showed

the MOSFET to have no advantage at Zgate under 100k ohms).



There's one voltage source that has very low ripple, good filtering, and

the same noise output as a low-value resistor. That's a thermopile.

Maybe the heatpump items aren't optimized for it, but one could

imagine using two-alloy printed wiring to get to some serious

potential differences in a compact, rugged, ultrareliable component.



You'd just have to allow a bit of warm-up time on that photon detector.

Cap multipliers are totally cool, when it comes to low noise.
If you haven't built one and tried it, you should.
I needed a low noise ~10 Watt supply. (30V, 0.3A)
I tried a couple of TIP31/ 32's.
a few (1-3 nV/rtHz.)*
(measured in 100Hz-100kHz band width)
Done.
George (don't argue with success) H.

*measured with an 8nV fet opamp

 

[whit3rd]

On Friday, May 30, 2014 6:59:20 AM UTC-7, Phil Hobbs wrote:

On 05/29/2014 01:44 PM, whit3rd wrote:

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output! You'd be differentially heating two (sets of) thermocouples,
and a common-mode temperature doesn't change the voltage at all.

 

[Phil Hobbs]

On 05/29/2014 01:44 PM, whit3rd wrote:

On Wednesday, May 28, 2014 7:27:09 PM UTC-7, Phil Hobbs wrote:
On 5/28/2014 4:20 PM, whit3rd wrote:

Have you considered direct shunt regulation?
[ AC-coupled example, with some DC stability issues unaddressed]

You're stuck with the wideband noise of the reference, though. I often
have to care about nanovolt 1-Hz noise on power supplies.

Classic solution: a string of mercury batteries. You can still do it, with
standard cells.

Horrible tempco, short lifetime, expensive, and completely unnecessary.
Some circuits require very low DC drift and noise in their power
supplies, but not many. Most of the time the low-frequency PSR of op
amps and current sources and such like will take care of the low
baseband instability.

The PSR of op amps falls off pretty badly with frequency, however, and
the noise current due to photodiode capacitance differentiating its bias
supply rises linearly. So you really want the supplies to be quiet at
high frequency, plus you want excellent rejection of SMPS ripple. Cap
multipliers are great for that, but normal voltage regulators aren't.
Thus the shunt-regulated cap multiplier is a big win.

Wideband noise in a '431 is likely recombination noise in the base currents,
as I understand it. So, maybe one with higher quiescent current would
be better. A FET solution (like current-limit diode for a reference and
jFET op amp for gain) would help, if things like popcorn/flicker didn't
take over.

Bandgaps are inherently noisy, because they have to put 20 dB of gain on
the delta-Vbe to get it to cancel out d(Vbe)/dT. Base current shot
noise (which is what I think you're talking about) is rarely dominant at
low current densities.

Even the humble follower transistor in the lightly-loaded C-multiplier, might
be noisier than a FET (or MOSFET,though I recall a discussion that showed
the MOSFET to have no advantage at Zgate under 100k ohms).

Your average low-sat transistor such as a ZXT11N15DFTA has about a
nanovolt 1-Hz noise at 1 mA of collector current. Your average bandgap
starts at 40 nV and goes up from there--a difference of 30 dB or more.

MOSFETs make horrible cap multipliers, because they're far noisier than
BJTs, their operating V_GS is much bigger than a V_BE drop, and they
have very low transconductance.

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.
Maybe the heatpump items aren't optimized for it, but one could
imagine using two-alloy printed wiring to get to some serious
potential differences in a compact, rugged, ultrareliable component.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

> You'd just have to allow a bit of warm-up time on that photon detector.

Cheers

Phil Hobbs


--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[Phil Hobbs]

On 5/30/2014 12:50 PM, whit3rd wrote:

On Friday, May 30, 2014 6:59:20 AM UTC-7, Phil Hobbs wrote:
On 05/29/2014 01:44 PM, whit3rd wrote:

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output! You'd be differentially heating two (sets of) thermocouples,
and a common-mode temperature doesn't change the voltage at all.

A thermopile is a bunch of thermocouples in series. Their output
voltage is approximately proportional to the temperature drop across the
junction pairs. That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT (3000
ppm/K at room temperature), and with feasible T_cold values, it's much
worse.

Cheers

Phil Hobbs





--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[whit3rd]

On Friday, May 30, 2014 2:21:01 PM UTC-7, Phil Hobbs wrote:

On 5/30/2014 12:50 PM, whit3rd wrote:

On Friday, May 30, 2014 6:59:20 AM UTC-7, Phil Hobbs wrote:

On 05/29/2014 01:44 PM, whit3rd wrote:



There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!

A thermopile is a bunch of thermocouples in series. Their output
voltage is approximately proportional to the temperature drop across the
junction pairs. That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER
proportional to absolute temperature in normal circumstances.

It's true, by the third law of thermodynamics, that thermocouple coefficients
(and other things) do vanish at absolute zero, but at room temperature a thermocouple
has no capability to give an absolute temperature reading, and that's
because the Seebeck equation contains no absolute temperature sensitivity.

 

[Phil Hobbs]

On 5/31/2014 2:49 AM, whit3rd wrote:

On Friday, May 30, 2014 2:21:01 PM UTC-7, Phil Hobbs wrote:
On 5/30/2014 12:50 PM, whit3rd wrote:

On Friday, May 30, 2014 6:59:20 AM UTC-7, Phil Hobbs wrote:

On 05/29/2014 01:44 PM, whit3rd wrote:



There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!


A thermopile is a bunch of thermocouples in series. Their output
voltage is approximately proportional to the temperature drop across the
junction pairs. That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER
proportional to absolute temperature in normal circumstances.

It's true, by the third law of thermodynamics, that thermocouple coefficients
(and other things) do vanish at absolute zero, but at room temperature a thermocouple
has no capability to give an absolute temperature reading, and that's
because the Seebeck equation contains no absolute temperature sensitivity.

You miss my point. The output voltage is proportional to the
temperature drop across the junction, and so the tempco goes as
1/(Thot-Tcold), which is huge--generally _much bigger_ than PTAT.
(Tcold being zero kelvin was a fictitious best case for your argument,
with zero physicsy stuff in it--see the linear approximation above.)

Thus there's absolutely no reason to go to all that trouble, when a
simple shunt regulated cap multiplier will do much better for under a
buck, with a 1-nanovolt noise floor.

Cheers

Phil Hobbs



--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[whit3rd]

On Saturday, May 31, 2014 8:32:07 AM UTC-7, Phil Hobbs wrote:

On 5/31/2014 2:49 AM, whit3rd wrote:

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

... its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!

A thermopile is a bunch of thermocouples in series. ... That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER

You miss my point. The output voltage is proportional to the
temperature drop across the junction, and so the tempco goes as
1/(Thot-Tcold), which is huge--generally _much bigger_ than PTAT.
(Tcold being zero kelvin was a fictitious best case for your argument,
with zero physicsy stuff in it--see the linear approximation above.)

Actually, you need the physicsy stuff to find a regime where PTAT is
the case. Take 100x100mm of flexible circuit board, put
quarter-millimeter wires with quarter-millimeter spacings all over it.
That's 200 thermocouples. Roll it into cigarette-size, glue a base
on one end, and a heater resistor on the other.
Insulate everything but the base.

Send enough current through the resistor to raise the hot end temperature
100 degrees above the ambient-temperature end (i.e. set the
power at the thermal conductance of the wires times 100 degrees).

Thot - Tambient = heat_input / thermal_conductance = 100

Now look at dependence on ambient temperature Ta

dV/dTa = d/dTa (K) * (Thot - Ta) + K * d/dTa ( Thot - Ta)

but we know that the sum Thot - Ta is set by the heat input, that sum is constant!

So the temperature dependence is PTAT whenever the thermocouple has
a Seebeck coefficient that is proportional to temperature. That happens
at/near absolute zero (third law of thermodynamics).

The principle is not at all impractical; accurate RF current meters
have worked this way for decades, inside the box where one rarely
sees it happening.

 

[Maynard A. Philbrook Jr.]

In article <8436b95b-c9fa-4b36-911a-4dd625c904b2@googlegroups.com>,
whit3rd@gmail.com says...

On Friday, May 30, 2014 2:21:01 PM UTC-7, Phil Hobbs wrote:
On 5/30/2014 12:50 PM, whit3rd wrote:

On Friday, May 30, 2014 6:59:20 AM UTC-7, Phil Hobbs wrote:

On 05/29/2014 01:44 PM, whit3rd wrote:



There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

Ultrareliable isn't how I'd describe it. Also its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!


A thermopile is a bunch of thermocouples in series. Their output
voltage is approximately proportional to the temperature drop across the
junction pairs. That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER
proportional to absolute temperature in normal circumstances.

It's true, by the third law of thermodynamics, that thermocouple coefficients
(and other things) do vanish at absolute zero, but at room temperature a thermocouple
has no capability to give an absolute temperature reading, and that's
because the Seebeck equation contains no absolute temperature sensitivity.

Which is why you need a cold junction reference to measure against.

Jamie

 

[whit3rd]

On Saturday, May 31, 2014 12:44:23 PM UTC-7, Phil Hobbs wrote:

... A very garden-variety voltage reference can
do 30 ppm/K, and good ones are below 5 ppm/K (e.g. the LT1021).

Ah, if only life were that simple. To get best errors from a biased
photodiode, you need the bias voltage NOT to be temperature
independent (because the diode capacitance, and stored charge, isn't).
There's a square root of T in the formula...

 

[whit3rd]

On Saturday, May 31, 2014 2:41:02 PM UTC-7, Phil Hobbs wrote:

On 5/31/2014 5:30 PM, whit3rd wrote:

On Saturday, May 31, 2014 12:44:23 PM UTC-7, Phil Hobbs wrote:


... A very garden-variety voltage reference can
do 30 ppm/K, and good ones are below 5 ppm/K (e.g. the LT1021).

Ah, if only life were that simple. To get best errors from a biased
photodiode, you need the bias voltage NOT to be temperature
independent (because the diode capacitance, and stored charge, isn't).

What formula are you talking about, exactly?

The discussion started with stabilizing a bias voltage for
a photodiode; the current from a photodiode is

I = Iphoto + Isat + d/dt(C * Vbias)

one usually ignores (or calibrates out) the constant Isat
That last term isn't just one error, it's the sum of two terms

dC/dt * Vbias + C * d/dt(Vbias)

Thus, in a temperature-varying-with-time case, change 't' time for 'T',
temperature, and optimum photocurrent measurement requires

0 = dC/dT * Vbias + C * d/dT(Vbias)

d/dT( Vbias) = -( dC/dT * Vbias) / C

It's not worth worrying about unless low-frequency response is important,
or unless the amplifier could saturate, because temperature is a slow-changing
variable.

Sze (_Physics of Semiconductor Devices_) gives a formula for abrupt-junction
diode capacitance that seems appropriate

C = [constant] * sqrt(Vbi - V - 2kT/q)

where Vbi is the builtin voltage, V is the bias...

The PD bias is there to reduce its capacitance (by as much as 7 times),
and therefore to reduce the high frequency noise of the TIA, which is

i_N = e_NTIA * 2 pi f C_diode .

As long as the PD bias doesn't have significant noise in the bandwidth
of interest, you don't care about small variations.

 

[Phil Hobbs]

On 5/31/2014 3:33 PM, whit3rd wrote:

On Saturday, May 31, 2014 8:32:07 AM UTC-7, Phil Hobbs wrote:
On 5/31/2014 2:49 AM, whit3rd wrote:

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

... its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!

A thermopile is a bunch of thermocouples in series. ... That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER

You miss my point. The output voltage is proportional to the
temperature drop across the junction, and so the tempco goes as
1/(Thot-Tcold), which is huge--generally _much bigger_ than PTAT.
(Tcold being zero kelvin was a fictitious best case for your argument,
with zero physicsy stuff in it--see the linear approximation above.)

Actually, you need the physicsy stuff to find a regime where PTAT is
the case. Take 100x100mm of flexible circuit board, put
quarter-millimeter wires with quarter-millimeter spacings all over it.
That's 200 thermocouples. Roll it into cigarette-size, glue a base
on one end, and a heater resistor on the other.
Insulate everything but the base.

Send enough current through the resistor to raise the hot end temperature
100 degrees above the ambient-temperature end (i.e. set the
power at the thermal conductance of the wires times 100 degrees).

Thot - Tambient = heat_input / thermal_conductance = 100

Now look at dependence on ambient temperature Ta

dV/dTa = d/dTa (K) * (Thot - Ta) + K * d/dTa ( Thot - Ta)

but we know that the sum Thot - Ta is set by the heat input, that sum is constant!

So the temperature dependence is PTAT whenever the thermocouple has
a Seebeck coefficient that is proportional to temperature. That happens
at/near absolute zero (third law of thermodynamics).

The principle is not at all impractical; accurate RF current meters
have worked this way for decades, inside the box where one rarely
sees it happening.

You're just blowing smoke. A very garden-variety voltage reference can
do 30 ppm/K, and good ones are below 5 ppm/K (e.g. the LT1021). Good
luck getting anywhere near that with a thermopile, especially one in
SOT-23.

Cheers

Phil Hobbs

--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[whit3rd]

On Saturday, May 31, 2014 4:41:21 PM UTC-7, Phil Hobbs wrote:
> On 5/31/2014 7:21 PM, whit3rd wrote:

[about error current in a photodiode]

dC/dt * Vbias + C * d/dt(Vbias)

There's no error term in the external circuit from dC/dt.

Of course there is! The charge on the capacitor doesn't
move through the depletion region, it goes in the external wiring.

The temperature of the photodiode changes on a timescale of minutes to
hours. If you have a capacitance of 10 pF at 25 C...
The current error resulting from your mechanism would be

delta I = (dT/dt)*dC/dT*V_bias

= (175 K)/(30s) * (1 pF/100 K) * 10 V = 0.6 pA.

How many sensitive photoreceivers have you actually designed?

One, in particular, was for fluorescence spectroscopy on microscopic
samples. It had a ~1 cm photodiode to cover the monochromator's exit
slit, and took a minute or so to scan the spectrum with a stepping motor
moving the grating. The output was picoamps. So was leakage. Capacitance
was high, seconds-to-minutes response was what worked best
with a chart recorder.

It mismatches by a few orders of magnitude the numbers you've plugged in.

 

[Phil Hobbs]

On 5/31/2014 5:30 PM, whit3rd wrote:

On Saturday, May 31, 2014 12:44:23 PM UTC-7, Phil Hobbs wrote:

... A very garden-variety voltage reference can
do 30 ppm/K, and good ones are below 5 ppm/K (e.g. the LT1021).

Ah, if only life were that simple. To get best errors from a biased
photodiode, you need the bias voltage NOT to be temperature
independent (because the diode capacitance, and stored charge, isn't).
There's a square root of T in the formula...

What formula are you talking about, exactly?

The PD bias is there to reduce its capacitance (by as much as 7 times),
and therefore to reduce the high frequency noise of the TIA, which is

i_N = e_NTIA * 2 pi f C_diode .

As long as the PD bias doesn't have significant noise in the bandwidth
of interest, you don't care about small variations.

At one point, we were talking about voltage regulators, and the noise
rejection thereof.

Cheers

Phil Hobbs

--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net

 

[John Larkin]

On Sat, 31 May 2014 12:33:52 -0700 (PDT), whit3rd <whit3rd@gmail.com> wrote:

On Saturday, May 31, 2014 8:32:07 AM UTC-7, Phil Hobbs wrote:
On 5/31/2014 2:49 AM, whit3rd wrote:

There's one voltage source that has very low ripple, good filtering, and
the same noise output as a low-value resistor. That's a thermopile.

... its tempco would be on
the order of 3000 ppm/K.

Not so. That's the tempco of the series resistance, not of the voltage
output!

A thermopile is a bunch of thermocouples in series. ... That means that V = K(T_hot -T_cold).

Even if T_cold = 0 kelvin, that scheme can't do better than PTAT

That is a tremendously improbable case! Thermopiles operated on
HEAT SOURCES stabilize to a fixed gradient (established by heat conduction)
and one expects (T_hot - T_cold) to be a constant. It is NEVER

You miss my point. The output voltage is proportional to the
temperature drop across the junction, and so the tempco goes as
1/(Thot-Tcold), which is huge--generally _much bigger_ than PTAT.
(Tcold being zero kelvin was a fictitious best case for your argument,
with zero physicsy stuff in it--see the linear approximation above.)

Actually, you need the physicsy stuff to find a regime where PTAT is
the case. Take 100x100mm of flexible circuit board, put
quarter-millimeter wires with quarter-millimeter spacings all over it.
That's 200 thermocouples. Roll it into cigarette-size, glue a base
on one end, and a heater resistor on the other.
Insulate everything but the base.

Send enough current through the resistor to raise the hot end temperature
100 degrees above the ambient-temperature end (i.e. set the
power at the thermal conductance of the wires times 100 degrees).

Thot - Tambient = heat_input / thermal_conductance = 100

Now look at dependence on ambient temperature Ta

dV/dTa = d/dTa (K) * (Thot - Ta) + K * d/dTa ( Thot - Ta)

but we know that the sum Thot - Ta is set by the heat input, that sum is constant!

So the temperature dependence is PTAT whenever the thermocouple has
a Seebeck coefficient that is proportional to temperature. That happens
at/near absolute zero (third law of thermodynamics).

The principle is not at all impractical; accurate RF current meters
have worked this way for decades, inside the box where one rarely
sees it happening.

My wife gave me this as a birthday present. Can't imagine where she got it.

https://dl.dropboxusercontent.com/u/53724080/Gear/HF_Ammeter.JPG




--

John Larkin Highland Technology Inc
www.highlandtechnology.com jlarkin at highlandtechnology dot com

Precision electronic instrumentation

 

[Phil Hobbs]

On 5/31/2014 7:21 PM, whit3rd wrote:

On Saturday, May 31, 2014 2:41:02 PM UTC-7, Phil Hobbs wrote:
On 5/31/2014 5:30 PM, whit3rd wrote:

On Saturday, May 31, 2014 12:44:23 PM UTC-7, Phil Hobbs wrote:


... A very garden-variety voltage reference can
do 30 ppm/K, and good ones are below 5 ppm/K (e.g. the LT1021).

Ah, if only life were that simple. To get best errors from a biased
photodiode, you need the bias voltage NOT to be temperature
independent (because the diode capacitance, and stored charge, isn't).

What formula are you talking about, exactly?

The discussion started with stabilizing a bias voltage for
a photodiode; the current from a photodiode is

I = Iphoto + Isat + d/dt(C * Vbias)

one usually ignores (or calibrates out) the constant Isat
That last term isn't just one error, it's the sum of two terms

dC/dt * Vbias + C * d/dt(Vbias)

There's no error term in the external circuit from dC/dt. It's caused
by the spreading of the shielding regions on the edges of the depletion
region, and doesn't have to be charged up from zero volts.

Thus, in a temperature-varying-with-time case, change 't' time for 'T',
temperature, and optimum photocurrent measurement requires

0 = dC/dT * Vbias + C * d/dT(Vbias)

d/dT( Vbias) = -( dC/dT * Vbias) / C

Have you ever actually designed a photoreceiver? Doesn't sound like it.
You're just blowing smoke.

It's not worth worrying about unless low-frequency response is important,
or unless the amplifier could saturate, because temperature is a slow-changing
variable.

Sze (_Physics of Semiconductor Devices_) gives a formula for abrupt-junction
diode capacitance that seems appropriate

C = [constant] * sqrt(Vbi - V - 2kT/q)

The temperature of the photodiode changes on a timescale of minutes to
hours. If you have a capacitance of 10 pF at 25 C, it might change by a
picofarad or so between there and 100 C. Typical parameters might be a
10V reverse bias, and a very generous estimate of the thermal TC for the
assembly might be 30 seconds.

For an extreme example, say you dunked the whole thing into boiling oil
at 200 C, i.e. a 175K transient.

The current error resulting from your mechanism would be

delta I = (dT/dt)*dC/dT*V_bias

= (175 K)/(30s) * (1 pF/100 K) * 10 V = 0.6 pA.

Not very impressive numbers, even for an extreme case, and of course it
would almost all be well below 1 Hz.

where Vbi is the builtin voltage, V is the bias...

The PD bias is there to reduce its capacitance (by as much as 7 times),
and therefore to reduce the high frequency noise of the TIA, which is

i_N = e_NTIA * 2 pi f C_diode .

As long as the PD bias doesn't have significant noise in the bandwidth
of interest, you don't care about small variations.

How many sensitive photoreceivers have you actually designed?

Cheers

Phil Hobbs

--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot nethttp://electrooptical.net

 

[Maynard A. Philbrook Jr.]

In article <qdkko95v4baiom25edjh6tr3497vcuklgg@4ax.com>,
jjlarkin@highNOTlandTHIStechnologyPART.com says...

The principle is not at all impractical; accurate RF current meters
have worked this way for decades, inside the box where one rarely
sees it happening.

My wife gave me this as a birthday present. Can't imagine where she got it.

https://dl.dropboxusercontent.com/u/53724080/Gear/HF_Ammeter.JPG


I haven't seen one of those around in years.

Jamie

 

[Phil Hobbs]

On 5/31/2014 8:28 PM, whit3rd wrote:

On Saturday, May 31, 2014 4:41:21 PM UTC-7, Phil Hobbs wrote:
On 5/31/2014 7:21 PM, whit3rd wrote:

[about error current in a photodiode]
dC/dt * Vbias + C * d/dt(Vbias)

There's no error term in the external circuit from dC/dt.

Of course there is! The charge on the capacitor doesn't
move through the depletion region, it goes in the external wiring.

So if you have an unconnected photodiode, what happens when its
temperature changes? You still have a depletion region, and its width
still changes with temperature.

The temperature of the photodiode changes on a timescale of minutes to
hours. If you have a capacitance of 10 pF at 25 C...
The current error resulting from your mechanism would be

delta I = (dT/dt)*dC/dT*V_bias

= (175 K)/(30s) * (1 pF/100 K) * 10 V = 0.6 pA.

How many sensitive photoreceivers have you actually designed?

One, in particular, was for fluorescence spectroscopy on microscopic
samples. It had a ~1 cm photodiode to cover the monochromator's exit
slit, and took a minute or so to scan the spectrum with a stepping motor
moving the grating. The output was picoamps. So was leakage. Capacitance
was high, seconds-to-minutes response was what worked best
with a chart recorder.

It mismatches by a few orders of magnitude the numbers you've plugged in.

Do tell.

Cheers

Phil Hobbs


--
Dr Philip C D Hobbs
Principal Consultant
ElectroOptical Innovations LLC
Optics, Electro-optics, Photonics, Analog Electronics

160 North State Road #203
Briarcliff Manor NY 10510

hobbs at electrooptical dot net
http://electrooptical.net